1. Field of the Invention
The present invention relates to switching regulators and particularly to controlling the charging switch of a switching regulator with an inductance, a filter capacitance and a diode.
2. Description of the Related Art
FIG. 8 shows a known down converter with a simple switch, such as, for example, shown in “Halbleiter-schaltungstechnik” U. Tietze, C. H. Schenk, Springer-Verlag, 9. Auflage, 1989, illustration 18.37 on page 564. The down converter in FIG. 8 comprises a ring-like circuit with a coil 400, a capacitor 402 and a diode 404. Further, the down converter or switching regulator, respectively, in FIG. 8 comprises a charging switch 406 as well as a control not shown in FIG. 8, which is formed to control the charging switch 406 such that the output voltage of the switching regulator, indicated by USR in FIG. 8, is maintained on a defined level or in a range around the defined level, respectively.
The circuit shown in FIG. 8 comprises a determined number of nodes, which will be discussed below. A pole of an input voltage source U0 is connected to a first input node 410 of the circuit, while another potential of the input source U0 is connected to a second input node 412. The second input node 412 is typically the ground node. A first output node 414 is also referred to as first output rail or positive output rail, while a second output node 416 is also referred to as second output rail or negative output rail, respectively, when the convention shown in FIG. 8 is used for the output voltage of the switching regulator USR. On the one hand, the switch 406 is connected between the first input node 410 and a first intermediate node 418. Further, the diode 404 is connected between the first intermediate node 418 and the second input node 412, such that the anode of the diode is connected to the second input node 412, while the cathode of the diode is connected to the first input node 418. Further, as shown in FIG. 8, the capacitor 402 is connected between the first output node 414 and the second output node 416. According to the configuration of the network of diode, coil and capacitor shown in FIG. 8, the coil is connected between the first intermediate node 418 and the first output node 414.
Below, the functionality of the circuit shown in FIG. 8 will be discussed. As long as the switch 406 is closed, UD is equal to the negative input voltage U0. If it opens, the inductor current IL maintains its direction, and the amount of UD sinks, until the diode becomes conductive, which means to about 0 potential. The time curve of the coil current results from the law of induction, according to which the voltage at the coil is equal to the product of inductance L of the coil and the derivation of the coil current according to time. During the switch-on time, which means when the input voltage U0 is applied to the diode 404, the voltage U0−USR is applied to the inductor. During the switch-off time taus of the switch 406, the voltage UL=−USR is applied to the inductor. Therefore, an amount of current change ΔIL results, which is as follows:ΔIL=1/L·(−USR)·Δtaus=1/L(U0−USR)teinFrom this balance, the output voltage can be calculated again, which is defined as follows:USR=[tein/(tein+taus)]·U0=tein/T·U0=p·U0.In the previous equation, T=tein·+taus=1/F is the oscillation period and p=tein/T is the so called duty cycle. As expected, it can be seen that the arithmetic average of UD results as output voltage. Typically, the inductance L of the coil 400 is chosen such that the minimum current is not undershot, as is known in the art. Further, it is known that by increasing the clock frequency, the inductance can be reduced. Further, with too high frequencies, the effort for the switching transistor and the control circuit increases. Additionally, dynamic switching losses increase in proportion to the frequency.
The capacitor 402 determines the ripple of the output voltage. Generating the switching signal for switching the charging transistor 406 is usually performed by a pulse width modulator and a regulator with voltage reference. In particular, a reference voltage providing a set value is supplied to a subtracter, to which the current output voltage USR is also supplied as actual value. The output signal of the subtracter is supplied to a variable gain amplifier, feeding a comparator, to which, on the other hand, a signal generated by a saw tooth generator is supplied. The output signal of the comparator is the control signal for the switch 406 in FIG. 8. The variable gain amplifier is typically a PI variable gain amplifier. The same increases its output signal for so long until the difference at the output of the subtracter becomes 0, which means until the output voltage USR is equal to the set output voltage. Typical ranges for dimensioning the coil are in the millihenry range (e.g. 2.7 mH), while typical values for capacitors are in the three-digit micro Farad range (e.g. 100 μF), when switching frequencies in the range of 50 kHz are used.
Switching regulators shown in FIG. 8 are to provide a suitable voltage supply to a subsequently connected circuit, such as an ASIC. The voltage supply consists normally of one or several constant direct voltages of, for example, +5 V or ±15 V. Frequently, the same is not available in the desired form from the start and has to be generated first by, for example, a switching regulator shown in FIG. 8 that can be supplemented by a downstream linear regulator to remove the ripples of the output voltage. Usually, a rectifier is at the input side of the switching regulator shown in FIG. 8 which generates the input voltage U0 from the alternating current or three-phase current net (230 V or 400 V), respectively, of the power station.
Thus, in deviation from the switching regulator shown in FIG. 8, other regulators exist, with a transformer, a rectifier, a smoothing capacitor and possibly a linear regulator for voltage stabilization. However, the transformer is difficult to produce and thus expensive. Further, it requires a lot of space. A further disadvantage of the transformer is its frequency-depending working range. This is, for example, limited to the network frequency of 50 Hz or 60 Hz, respectively. If the frequency deviates, this also causes a deviation of the output voltage of the transformer. The voltage transfer does not work with a direct voltage at the input.
If the transformer is omitted and only rectifier, smoothing capacitor and linear regulator are used, a lot of energy in the form of heat is lost. Additionally, sufficient cooling of the linear regulator has to be provided, which is again very expensive and requires space. All this is avoided by using a switching regulator instead of the linear regulator, as illustrated with regard to FIG. 8. By the significantly better efficiency, little energy in the form of heat is lost and thus the effort for the cooling is significantly lower. As has already been discussed, the switching regulator requires an inductor (the inductance 400 in FIG. 8), which is relatively expensive in the production. However, the same has only one winding and is thus simpler to produce than a transformer having two windings. Above that, the inductor can be minimized by selecting a higher operating frequency, which also works for transformers.
Many known switching network parts, such as the switching network part shown in FIG. 8, are problematical in some regard. Normally, the input voltage range is limited to a ratio of UE,max/UE,min≦5, which can be seen from catalogs of different providers. This range is too low for some applications and should be increased to a ratio of about 20:1.
The voltage supply of the regulator itself is either performed via a separate voltage source or is generated from the input voltage, which means an additional voltage regulator and thus additional effort.
Further, for a flexible usage, it is intended to be able to select the input voltage significantly higher than the maximum allowable operating voltage of the regulator itself, without using additional voltage regulators for generating this operating voltage.
Additionally, when applying the input voltage, a possibly fast controlled starting of the switching regulator should be ensured. This so called starting-delay should be as small as possible, particularly for time-critical applications.
As has already been discussed, the switching regulator of the down converter type shown in FIG. 8, which is also referred to as buck converter in the art, is based on, the fact that the charging switch 406 is switched on to charge the positive output node 414 against the negative output node 416, and then switch the switch off again, which means to open it, so that the output voltage does not exceed a set value range determined by the average of the finally resulting output voltage USR. Possibilities of regulating the switch are basically filtering the output voltage to obtain an average of the output voltage without ripples, and then controlling the charging switch based on the filtered output voltage by a set/actual comparison. Alternatively, there is also the possibility to regulate the switching regulator based on the current flowing through the coil 400, since the current flowing through the coil 400 is connected to the output voltage, since the output voltage is controlled by charging the node 414 of the capacitor 402. Thus, current regulators exist, which detect the current through the coil 400, filter the current value to obtain an average current value, and which then control the charging switch 406 based on the set/actual comparison due to the filtered current average.
Different regulation concepts are described in DE 19814681 A1, DE 19505417 A1, DE 19933039, EP 0759653 or EP 0664596. Thus, DE 19814681 A1 shows, for example, a current mode switching regulator with a first regulating means for voltage regulation and a second regulation means for load current regulation, each having two inputs and one output, wherein a reference signal can be supplied to a first input of the first regulation means and an output signal of the current mode switching regulator to a second input as a variable, and wherein the output of the first regulating means is coupled to a first input of the second regulator means. Further, a power switch controllable by a control signal of the second regulating means is provided, whose load path is disposed between a first pole with a first supply potential and a second pole with a second supply potential. An inductor means is disposed in series to this load path. Further, an integrator is provided, which generates a regulating signal sampling a load current by time integration of the inductor voltage falling at the inductor means, which is coupled into a second input of the second regulating means as variable. The voltage falling at the inductor inductance is integrated over time by the integrator to detect the current through the inductor.
DE 19933039 A1 discloses an apparatus for generating a regulating signal for a direct voltage converter. The apparatus for generating the regulating signal comprises a voltage regulator and a current regulator. A limiter is provided between the voltage regulator and the current regulator, which serves for limiting the output signal of the voltage regulator. The limiter comprises an input terminal, across which a determinable limit signal for the current set value can be input.
It is a problem of set value regulators operating with a fixed set value, that this set value has to be compared with an actual value. Normally, the set value exists as fixed value, which means as direct signal, while the actual value constantly varies fast, which is an inherent property of the switching regulator, since the ripples of the output voltage as well as the ripples of the output current occur systematically by opening and closing the charging switch. Thus, in these known regulator concepts, an actual value of either the voltage or the current has to be filtered, to obtain a filtered actual value having a direct voltage characteristic, such that a set/actual comparison can be performed at all with the set value present in the “direct current characteristic”.
By this filtering, an additional time constant is introduced in the locked loop. Additionally, there is the problem that the current locked loop becomes unstable with higher pulse width ratios. Typically, known current locked loops run only stable up to a pulse width ratio of 50%. With larger pulse width ratios, such a switching regulator becomes instable. This means that a change of the output voltage can no longer be reasonably compensated. For avoiding this instability already at the pulse width ratio of 50%, which means for increasing the pulse width ratio range across 50%, normally, circuits are provided, for example circuits overlaying a saw tooth-shaped voltage across the current limit value predetermined by the voltage regulator to countereffect this instability effect. Thereby, a current locked loop can also be stabilized for pulse ratios of more than 50%. However, this means increased switching effort, since the saw tooth curve has to be generated and laid over the current set value. Further, the switching effort for the filter for filtering the current actual value curve has to be considered. Thus, known concepts have a double disadvantage, since they require a filter for filtering the current value curve, and further require a stabilizing circuit, partly caused by this filter, to realize pulse width ratios of more than 50%.
These additional switching parts have the effect that switching regulators becomes more expensive, particularly when they are to be designed integrably, which is not least caused by the fact that any additional component in an integrated circuit increases the rejection probability for the whole integrated circuit.
Even when such a known switching regulator is not designed in integrated design, the additional switching parts still have to be designed and realized, which can cause cost increases.